Universal pulse width modulating power converter

ABSTRACT

A controller for a pulse width modulating (PWM) power converter. The controller monitors an output voltage and a current through a magnetic element for modulating a duty cycle of a main power switch. The controller is an eight-pin integrated circuit which controls either a forward converter or a post regulator without internal modifications. The monitored current of the forward converter forms a positive sensing signal. The monitored current of the post regulator forms a negative sensing signal. A current sense circuit of the controller forms an absolute value of either sensing signal. A soft-start circuit of the controller forms a start-up voltage ramp which is proportional to the level of a V CC  supply without requiring an external capacitor. A duty cycle of the main switch is gradually increased by comparing the start-up voltage ramp to the absolute value of the sensing signal. A pulse skipping circuit of the controller disables switching of the main switch under light load conditions. An error signal representative of a difference between the output voltage and a desired output voltage is compared to a pulse skip reference voltage. When the error signal falls below the reference voltage, the main switch is disabled. Because the pulse skip reference voltage is inversely related to the V CC  supply, pulsing of the main switch under light load conditions is spread out in time. A clock signal utilized to control switching of the main switch is generated internally or externally to the integrated circuit.

RELATED APPLICATION

This application is a continuation-in-part of U.S. patent application Ser. No. 09/231,523, filed, Jan. 24, 1999, and entitled, “INTERLEAVED ZERO CURRENT SWITCHING IN A POWER FACTOR CORRECTION BOOST CONVERTER”, now U.S. Pat. No. 6,091,233, the contents of which are hereby incorporated by reference.

FIELD OF THE INVENTION

The invention relates to the field of switching electrical power converters. More particularly, the invention relates to the field of pulse width modulating forward converters and post regulators.

BACKGROUND OF THE INVENTION

FIG. 1 illustrates a schematic block diagram of a conventional pulse width modulating (PWM) forward converter. As illustrated in FIG. 1, a voltage source V_(S) is coupled to a first terminal of a primary winding of a transformer T. A second terminal of the primary winding of the transformer T is coupled to a drain of a MOS transistor M. A source of the transistor M is coupled to a first terminal of a resistor R_(SENSE). A second terminal of the resistor R_(SENSE) is coupled to a first ground node. A voltage signal I_(SENSE) formed at the first terminal of the resistor R_(SENSE) is representative of a level of current passing through the primary winding of the transformer T when the transistor M is active.

A first terminal of a secondary winding of the transformer T is coupled to an anode of a diode D. A cathode of the diode D is coupled to a first terminal of a capacitor C₁, to an output node, and to a first terminal of a resistor R₁. A second terminal of the secondary winding of the transformer T and a second terminal of the capacitor C₁ are coupled to a second ground node. A second terminal of the resistor R₁ is coupled to a first terminal of a resistor R₂ and to an inverting input of an error amplifier A. A second terminal of the resistor R₂ is coupled to the second ground node. The resistors R₁ and R₂ form a resistive divider for supplying the amplifier A with a signal which is proportional to an output voltage V_(OUT) at the output node. The amplifier A can include optical elements so as to optically isolate the first ground node from the second ground node.

A non-inverting input of the amplifier A is coupled to a reference voltage V_(REF). The reference voltage V_(REF) is representative of a desired level for the output voltage V_(OUT). An output of the amplifier A forms an error signal V_(EA) and is coupled to a non-inverting input of a comparator CMP₁. The error signal V_(EA) is representative of a difference between the output voltage V_(OUT) and a desired level for the output voltage. The first terminal of the resistor R_(SENSE) is coupled to an inverting input of the comparator CMP₁ and to an inverting input of a comparator CMP₂. An output of the comparator CMP₁ is coupled to a first input of a logic NAND gate U₁. A current source is coupled to a first terminal of a capacitor C₂ and to a non-inverting input of the comparator CMP₂. A second terminal of the capacitor C₂ is coupled to the first ground node. A voltage signal V_(START) is formed at the first terminal of the capacitor C₂.

An output of the comparator CMP₂ is coupled to a second input of the NAND gate U₁. An output of the NAND gate U₁ is coupled to a set input S of a flip-flop U₂. A reset input R of the flip-flop U₂ is coupled to receive a clock signal V_(CLK). An inverted output {overscore (Q)} of the flip-flop U₂ is coupled to a gate of the transistor M.

When the transistor M is active (turned on), current flows from the source V_(S) and through the primary winding of the transformer T. This stores energy as an electromagnetic field associated with the primary winding of the transformer T. When the transistor M is inactive (turned off), the electromagnetic field collapses. By turning the transistor M on and off, energy is transferred to the secondary winding of the transformer T which induces a current to flow in the secondary winding. The current in the secondary winding of the transformer T is rectified by the diode D so as to form a voltage across the capacitor C₁. A duty cycle utilized for operating the transistor M controls the level of the output voltage V_(OUT) formed at the output node.

FIGS. 2a-b illustrate timing diagrams for the signals V_(EA), I_(SENSE) and V_(CLK) of the PWM forward converter illustrated in FIG. 1. When the clock signal V_(CLK) transitions from a logical low voltage to a logical high voltage, the output {overscore (Q)} of the flip-flop U₂ transitions to a logic high voltage. This turns on the transistor M. Under these conditions, current flows through the transistor M and the resistor R_(SENSE), as illustrated in FIG. 2a by the signal I_(SENSE) ramping up. When the signal I_(SENSE) reaches the level of the error signal V_(EA), this causes the output of the comparator CMP₁ to change from a logic high voltage to a logic low voltage. As a result, the output of the NAND gate U₁ changes from a logic low voltage to a logic high voltage and the output {overscore (Q)} of the flip-flop U₂ transitions from a logic high voltage to a logic low voltage. This turns off the transistor M. Upon a next transition of the clock signal V_(CLK), this cycle repeats. Note that as the error signal V_(EA) increases, the transistor M stays on for a longer portion of each cycle of the clock signal V_(CLK) because more time is required for the signal I_(SENSE) to exceed the error signal V_(EA). Conversely, as the error signal V_(EA) falls, the transistor M stays on a smaller portion of each cycle of the clock signal V_(CLK) because less time is required for the error signal I_(SENSE) to exceed the error signal V_(EA). Accordingly, the output voltage at the node V_(OUT) is regulated to the desired level by adjusting the duty cycle of the transistor M according to requirements of a load (not shown) which can be coupled to the output node to receive the output voltage V_(OUT).

Under normal operating conditions, the voltage V_(START) is at a higher level than the error signal V_(EA). Accordingly, the output of the comparator CMP₂ is a logic high voltage when the output of the comparator CMP₁ changes. Therefore, under normal operating conditions, the output of the comparator CMP₂ does not affect the duty cycle of the transistor M and the PWM converter operates as described above.

Upon start up, however, the output voltage V_(OUT) is low. As a result, the error signal V_(EA) is relatively large. In absence of soft-start circuit elements, including the current source I, the capacitor C₂ and the comparator CMP₂, this large error signal would result in the transistor M being held on for a large portion of each cycle of the clock signal V_(CLK) while the forward converter attempted to rapidly increase the output voltage to the desired level. As a result, excessive current would flow through the transistor M which would tend to cause premature failure of the transistor M.

Instead, upon start up, the current source I is turned on and the signal V_(START) slowly ramps up. Before the level of the signal V_(START) exceeds the level of the signal V_(EA), the duty cycle of the transistor M is not influenced by the signal V_(EA), but by the signal V_(START). As a result, the duty cycle of the transistor M gradually increases until the level of the signal V_(START) exceeds the level of the error signal V_(EA).

While the soft-start circuit elements of FIG. 1 provide a useful function, they also result in a disadvantage, especially when elements of the forward converter are incorporated into an integrated circuit. More particularly, so that the signal V_(START) ramps up slowly, the current produced by the current source I must be small in relation to the size of the capacitor C₂. This constraint either requires that the capacitor C₂ be external to the integrated circuit, which increases the pin count of the integrated circuit and, thus, the cost of producing the integrated circuit, or requires that the current produced by the current source I be so small as to be easily overwhelmed by noise and other transient signals, which reduces reliability.

Therefore, what is needed is improved soft-start technique for a PWM power converter.

Further, prior integrated circuits for controlling PWM power converters have been specifically tailored to the intended application. For example, a different integrated circuit design is utilized for a PWM forward converter than is utilized for a PWM post-regulator. This requirement of multiple integrated circuit designs tends to increase the costs associated with each.

Therefore, what is needed is a universal integrated circuit for controlling a PWM power converter.

SUMMARY OF THE INVENTION

The invention is a universal controller for a pulse width modulating (PWM) power converter. The controller monitors an output voltage of the power converter and a current through a magnetic element of the power converter for modulating a duty cycle of a main power switch of the power converter. The main power switch is closed in response to a transition in a clock signal. When the main power switch is closed, the current through the magnetic element forms a sensing signal (current ramp) representative of the current through the main power switch. The sensing signal is compared to an error signal representative of a difference between the output voltage and a desired level for the output voltage. When the sensing signal exceeds the error signal, the main power switch is opened. Opening and closing of the main power switch draws power from an input voltage source for forming the output voltage. In this manner, the duty cycle of the main power switch is controlled in a feedback loop. In a preferred embodiment, the controller is implemented as an eight pin integrated circuit.

According to an aspect of the present invention, the controller can be utilized for a power converter which is either a PWM forward converter or a PWM post regulator where differences between the PWM forward converter and the PWM post regulator are exclusively in circuitry external to the controller. In particular, the magnetic element of the power converter is a transformer; the main power switch for the PWM forward converter controls a current through a primary side of the transformer, whereas, the main power switch for the PWM post regulator controls a current through the secondary side of the transformer. When the power converter is a PWM forward converter, the sensing signal is positive in polarity, whereas, when the power converter is a PWM post regulator, the sensing signal is negative is polarity. A current sense circuit included in the controller forms a signal which is representative of the absolute value of the sensing signal for comparison to the error signal.

According to another aspect of the present invention, the controller includes a soft-start circuit which gradually increases a duty cycle of the main power switch upon start-up of the power converter. The soft-start circuit monitors the ramping up of a V_(CC) power supply and, in response, forms a start-up voltage ramp. The start-up voltage ramp begins ramping when the V_(CC) power supply reaches a first predetermined voltage level and is substantially proportional to a level of the V_(CC) power supply as the level of V_(CC) exceeds the first predetermined voltage level. In a preferred embodiment, the start-up voltage ramp is formed by generating a current which is substantially proportional to the level of the V_(CC) supply (minus the first predetermined voltage level) and by applying this current to a resistor, such that the start-up voltage ramp is formed across the resistor. During start-up, the start-up voltage ramp is compared to the current ramp for controlling the duty cycle of the main power switch. As a result, the duty cycle gradually increases upon start-up as the level of the voltage supply increases. Unlike prior arrangements, the start-up circuit does not require an external capacitor for forming the start-up voltage ramp. This reduces the number pins required when the controller as implemented as an integrated circuit.

According to a further aspect of the present invention, a pulse skipping circuit disables switching of the main power switch when a load powered by the power converter draws a low level of current. When the output voltage rises, as tends to occur when the load draws a low level of current, the error signal decreases. The error signal is compared to a pulse skip reference voltage. When the level of the error signal falls below the level of the pulse skip reference voltage, the main power switch is disabled until the error signal rises again. Preferably, the pulse skip reference voltage is inversely related to the supply voltage V_(CC). Accordingly, when the supply voltage V_(CC) is at a higher level, the output voltage must rise to a higher level before the main power switch is disabled than when the supply voltage V_(CC) is at a lower level. Therefore, forming the pulse skip reference voltage such that it is inversely related to the supply voltage tends to aid in spreading out in time pulsing of the main power switch under light load conditions. This tends to reduce switching noise while increasing efficiency.

According to yet another aspect of the present invention, the clock signal which is utilized to control switching of the main power switch can be selectively generated internally to the integrated circuit or externally to the integrated circuit. It is expected that when the power converter is a PWM forward converter, the clock signal is internally generated, whereas, when the power converter is a PWM post regulator, the clock signal is externally generated for synchronizing switching of the PWM post regulator with that of a pre-regulator.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates a schematic block diagram of a conventional pulse width modulating (PWM) forward converter.

FIGS. 2a-b illustrate timing diagrams for signals of the PWM forward converter illustrated in FIG. 1

FIG. 3 illustrates a schematic block diagram of a universal integrated circuit controller for a PWM power converter in accordance with the present invention.

FIG. 4 illustrates a schematic diagram of a PWM forward converter which incorporates the integrated circuit controller illustrated in FIG. 3.

FIGS. 5a-h illustrate timing diagrams for selected signals of the controller and PWM forward converter illustrated in FIGS. 3-4.

FIG. 6 illustrates a schematic block diagram of the soft-start circuit illustrated in FIG. 3.

FIGS. 7a-b illustrate a timing diagrams for selected signals of the soft-start circuit illustrated in FIGS. 3 and 6.

FIG. 8 illustrates a more detailed schematic diagram of the soft-start circuit illustrated in FIGS. 3 and 6.

FIG. 9 illustrates a schematic diagram of the PWM comparator illustrated in FIG. 3.

FIG. 10 illustrates a schematic block diagram of the pulse skip circuit illustrated in FIG. 3.

FIG. 11 illustrates a more detailed schematic diagram of the pulse skip circuit illustrated in FIGS. 3 and 10.

FIG. 12 illustrates a block schematic diagram of the current limit circuit illustrated in FIG. 3.

FIG. 13 illustrates a schematic diagram of a PWM post regulator which incorporates the integrated circuit controller illustrated in FIG. 3.

FIGS. 14a-e illustrate timing diagrams for selected signals of the post regulator illustrated in FIG. 13.

FIG. 15 illustrates a block schematic diagram of the current sense circuit illustrated in FIG. 3.

FIG. 16 illustrates a power converter circuit in accordance with the present invention including multiple post-regulators coupled to respective windings of a single transformer.

DETAILED DESCRIPTION OF A PREFERRED EMBODIMENT

FIG. 3 illustrates a schematic block diagram of a universal controller 100 for a pulse width modulating (PWM) power converter in accordance with the present invention. In the preferred embodiment, the controller 100 is implemented as an integrated circuit which includes eight pins, numbered 1-8 in FIG. 3. Pin 1 receives a signal GM from circuit elements external to the controller 100 and is coupled to an inverting input of an error amplifier A₁. Preferably, the amplifier A₁ is a transconductance (gm) amplifier. A non-inverting input of the amplifier A₁ is coupled to a reference voltage V_(REF1). An output of the amplifier A₁ forms an error signal V_(EA1) and is coupled to a first non-inverting input of a three-input PWM comparator CMP₃, to Pin 2 and to a first input of a pulse skip circuit 102. Pin 2 receives a signal V_(DC) from circuit elements external to the controller 100 when the controller 100 is utilized in a forward converter. However, Pin 2 is coupled to compensation elements when the controller 100 is utilized in a post-regulator, as explained in more detail herein. The signals V_(EA1) and V_(DC) can be present at the node coupled to the first non-inverting input of the comparator CMP₃. Only one of the signals V_(EA1), or V_(DC)actively controls a level of the voltage at this node at a time depending upon the particular application. An output of the pulse skip circuit 102 is coupled to a first input of a logic NOR gate U₃.

Pin 3 receives a voltage signal I_(SENSE1) or a voltage signal I_(SENSE2) from circuit elements external to the controller 100 and is coupled to an input of a current sense circuit 104 and to an input of a current limit circuit 106. An output of the current sense circuit 104 is a signal I_(SENX) which is coupled to an inverting input of the comparator CMP₃. The signal I_(SENX) applied to the inverting input of the comparator CMP₃ is representative of the signal I_(SENSE1) or the signal I_(SENSE2) which is applied to the input of the current sense circuit 104 though the polarity of the signal I_(SENX) is preferably positive regardless of the polarity of the signal I_(SENSE1) or I_(SENSE2) output of the current limit circuit 106 forms a signal I_(LIM) and is coupled to a second input of the NOR gate U₃.

Pin 4 receives a supply voltage V_(CC) from circuit elements external to the controller 100 and is coupled to a second input of the pulse skip circuit 102, to a first input of a soft-start circuit 108, to an inverting input of an hysteretic comparator CMP₄ and to a non-inverting input of an hysteretic comparator CMP₅. An inverting input of the comparator CMP₅ is coupled to a reference voltage V_(REF2). An output of the comparator CMP₅ forms a signal UVLO and is coupled to a second input of the soft-start circuit 108. An output of the soft-start circuit 108 forms a signal V_(START1) and is coupled to a second non-inverting input of the comparator CMP₃. An output of the comparator CMP₃ is coupled to a first input of a logic OR gate U₄.

Pin 5 is coupled to receive a signal SELECT from circuit elements external to the controller 100 and is coupled to an input of a reference adjust circuit 110. The reference adjust circuit 110 adjusts internal reference voltages for use by the controller 100 according to a logic level of the SELECT signal.

Pin 6 can be coupled to receive a signal SYNC from circuit elements external to the controller 100 and is coupled to an input of an oscillator 112. An output of the oscillator 112 forms a clock signal V_(CLK1) and is coupled to a second input of the OR gate U₄, to a reset input R of a reset-dominant flip-flop U₅ and to a first input of a logic NOR gate U₆. A clock signal generated externally to the controller 100 can be coupled to Pin 6 of the controller 100. In which case, the clock signal V_(CLK1) is synchronous with the externally generated clock signal. Otherwise, if an externally generated clock signal is not applied to Pin 6 of the controller 100, then the clock signal V_(CLK1) is generated by the oscillator 112 as a “free running” signal (i.e. the clock signal V_(CLK1) is not synchronous with any externally generated clock signal).

An output of the OR gate U₄ is coupled to a first input of a logic NAND gate U₇. An output of the NOR gate U₃ is coupled to a second input of the NAND gate U₇. An output of the NAND gate U₇ is coupled to a set input S of the flip-flop U₅. A Q output of the flip-flop U₅ is coupled to a second input of the NOR gate U₆. An output of the NOR gate U₆ is coupled to a first input of a logic AND gate U₈. An output of the comparator CMP₄ forms a signal OVP and is coupled to a second input of the AND gate U₈. An output of the AND gate U₈ is coupled to Pin 7 of the controller 100 and forms a switch control signal PWM OUT which is provided to circuit elements external to the controller 100.

FIG. 4 illustrates a schematic diagram of a PWM forward converter 200 which incorporates the controller 100 illustrated in FIG. 3. A voltage source V_(S) is coupled to a first terminal of a resistor R₃ and to a first terminal of a primary winding of a transformer T₁. A second terminal of the primary winding of the transformer T₁ is coupled to a drain of a MOS transistor M₁. Pin 7 of the controller is coupled to a gate of the transistor M₁. A source of the transistor M₁ is coupled to a first terminal of a sensing resistor R_(SENSE1) and to Pin 3 of the integrated circuit controller 100. A second terminal of the resistor R_(SENSE1) is coupled to a first ground node. The signal I_(SENSE1) is formed at the first terminal of the resistor R_(SENSE1) and is representative of a level of current passing through the primary winding of the transformer T₁ when the transistor M₁ is active (turned on). When the transistor M₁ is inactive (turned off), current flowing in the primary winding of the transformer T₁ can be returned to the supply V_(S) through a diode (not shown for simplicity of illustration).

A first terminal of a secondary winding of the transformer T₁ is coupled to an anode of a diode D₁. A cathode of the diode D₁ is coupled to a cathode of a diode D₂ and to a first terminal of an inductor L₁. A second terminal of the inductor L₁ is coupled to a first terminal of a capacitor C₃, to a first output node, and to a first terminal of an optical isolator 202. A second terminal of the secondary winding of the transformer T₁, an anode of the diode D₂ and a second terminal of the capacitor C₃ are coupled to a second ground node. An output voltage V_(OUT1) is formed at the first output node. A load 204 can be coupled across the capacitor C₃.

A second terminal of the optical isolator 202 forms the signal V_(DC) and is coupled to Pin 2 of the controller 100. The gm amplifier A₁ with its Pin 1 input tied to ground is utilized to supply a pull-up current for the optical isolator 202. A third terminal of the optical isolator 202 is coupled to a first terminal of a resistor R₄. A second terminal of the resistor R₄ is coupled to the first ground node. The signal V_(DC) is an error signal which is representative of a difference between the output voltage V_(OUT1) formed at the output node and a desired level for the output voltage V_(OUT1). A value of the resistor R₄ sets the desired level of the output voltage V_(OUT1). Because the optical isolator 202 forms the error signal V_(DC), the signal V_(DC) is applied to the first non-inverting input of the comparator CMP₃ (FIG. 3) and the amplifier A₁ (FIG. 3) is disabled from forming the error signal V_(EA1) . However, as mentioned above, the amplifier A₁ forms the pull-up current for the optical isolator 202. The optical isolator 202 optically isolates the first ground node from the second ground node. For example, the optical isolator 202 can include a TL431 optical isolator.

A first terminal of a second secondary winding of the transformer T₁ is coupled to an anode of a diode D₃. A cathode of the diode D₃ is coupled to a first terminal of a capacitor C₄, to a second output node, to Pin 4 of the controller 100 and to a second terminal of the resistor R₃. A supply voltage V_(CC) for the integrated circuit controller 100 is formed at the second output node. A second terminal of the second secondary winding of the transformer T₁, a second terminal of the capacitor C₄, Pin 5 of the controller 100 and Pin 8 of the controller 100 are coupled to the first ground node.

FIGS. 5a-h illustrate timing diagrams for selected signals of the controller 100 and PWM forward converter 200 illustrated in FIGS. 3-4. More particularly, FIG. 5a illustrates the clock signal V_(CLK1); FIG. 5b illustrates a signal at the Q output of the flip-flop U₅; FIG. 5c illustrates a signal at the output of the NOR gate U₆; FIG. 5d illustrates the switch control signal PWM OUT at Pin 7 of the controller 100; FIG. 5e illustrates a signal at the output of the OR gate U₄; FIG. 5f illustrates a signal at the output of the NAND gate U₇; FIG. 5g illustrates a signal at the output of the comparator CMP₃; and FIG. 5h illustrates the error signal V_(DC) and sensing signal I_(SENSE1).

Referring to FIGS. 3-5, upon the clock signal V_(CLK1) transitioning to a logic high voltage, as shown at time zero in FIG. 5a, the flip-flop U₅ (FIG. 3) is placed in a reset condition and, thus, the Q output of the flip-flop U₅ is a logic low voltage, as shown in FIG. 5b. Because the flip-flop U₅ is reset dominant, the flip-flop U₅ remains in a reset condition while the signal V_(CLK1) is a logic high voltage. In addition, the logic high voltage of the clock signal V_(CLK1) is applied to the first input of the NOR gate U₆ (FIG. 3). Under these conditions, the output of the NOR gate U₆ is a logic low voltage, as shown in FIG. 5c. Accordingly, the signal PWM OUT is a logic low voltage, as shown in FIG. 5d, and the transistor M₁ (FIG. 4) is held off while the clock signal V_(CLK1) is logic high voltage.

The logic high voltage of the clock signal V_(CLK1) is also applied to the second input of the OR gate U₄ (FIG. 3). As a result, the output of the OR gate U₄ is a logic high voltage, as shown in FIG. 5e. Assuming the signals P_(SKIP) and I_(LIM) are both a logic low voltage, then the output of the OR gate U₃ (FIG. 3) is logic high voltage. As a result, the output of the NAND gate U₇ (FIG. 3) and the set input S of the flip-flop U₅ are a logic low voltage, as shown in FIG. 5f.

Because the transistor M₁ is held off while the clock signal V_(CLK1) is a logic high voltage, the voltage signal I_(SENSE1) is lower than the error signal V_(DC), as shown in FIG. 5h. As a result, the output of the comparator CMP₃ (FIG. 3) is logic high voltage prior to the clock signal V_(CLK1) transitioning to a logic low voltage, as shown in FIG. 5g.

Then, when the clock signal V_(CLK1) transitions to a logic low voltage, as shown in FIG. 5a, the reset input of the flip-flop U₅ transitions to a logic low voltage. Thus, the flip-flop U₅ is no longer held in the reset condition. In addition, the logic high voltage of the output of the comparator CMP₃ results in the output of the OR gate U₄ remaining a logic high voltage upon the logic low voltage of the clock signal V_(CLK1) being applied to the second input of the OR gate U₄, as shown in FIG. 5e. As a result, the output of the NAND gate U₇ and the set input S of the flip-flop U₅ remain a logic low voltage upon a transition in the clock signal V_(CLK1) from high to low, as shown in FIG. 5f.

Because the set input S of the flip-flop U₅ is a logic low voltage, the Q output of the flip-flop U₅ remains a logic low voltage, as shown in FIG. 5b, and, thus, the second input of the NOR gate U₆ is also logic low voltage. In addition, the logic low voltage of the clock signal V_(CLK1) is applied to the first input of the NOR gate U₆. In response, the output of the NOR gate U₆ transitions to a logic high voltage, as shown in FIG. 5c. Accordingly, the signal PWM OUT also transitions to a logic high voltage, as shown in FIG. 5d.

As a result, the transistor M₁ is turned on. As current flows through the transistor M₁, the signal I_(SENSE1) begins to rise, as shown in FIG. 5h. Once the signal I_(SENSE1) rises above the level of the error signal V_(DC), then the output of the comparator CMP₃ transitions to a logic low voltage, as shown in FIG. 5g. As a result, the output of the OR gate U₄ transitions to a logic low voltage, as shown in FIG. 5e. Because the output of the OR gate U₄ is a logic low voltage, the output of the NAND gate U₇ transitions from a logic low voltage to a logic high voltage, as shown in FIG. 5f. This sets the flip-flop U₇, such that the Q output transitions to a logic high voltage, as shown in FIG. 5b. As a result, the output of the NOR gate U₆ and, thus, the signal PWM OUT become a logic low voltage, as shown in FIGS. 5c-d. This turns the transistor M₁ off. This process repeats for each cycle of the clock signal V_(CLK1), thereby regulating the output voltage V_(OUT1).

The above-described regulation of the output voltage V_(OUT1) can be altered or interrupted under certain conditions. For example, in the event of a fault condition which results in an excessive level for the output voltage supply V_(CC) (an overvoltage condition), the signal OVP transitions from a logic high voltage to a logic low voltage. In response, the AND gate U₈ holds the transistor M₁ off, thereby disabling switching of the transistor M₁.

In addition, under start-up conditions, the soft-start circuit 108 and the comparator CMP₃ illustrated in FIG. 3 alter operation of the controller 100. FIG. 6 illustrates a schematic block diagram of the soft-start circuit 108. The supply voltage V_(CC) is coupled to a non-inverting input of a comparator CMP₆, to a first terminal of a current source I₁ and to a first terminal of a current source I₂. An inverting input of the comparator CMP₆ is coupled to a reference voltage V_(REF4). A second terminal of the current source I₁ is coupled to a drain of a MOS transistor M₂. A gate of the transistor M₂ is coupled to receive the signal UVLO from the comparator CMP₅ (FIG. 3). A second terminal of the current source I₂ is coupled to a drain of a MOS transistor M₃. An output of the comparator CMP₆ is coupled to a gate of the transistor M₃. A source of the transistor M₂ and a source of the transistor M₃ are coupled to a first terminal of a resistor R₅. A second terminal of the resistor R₅ is coupled to the first ground node. The signal V_(START1) is formed across the resistor R₅.

FIGS. 7a-b illustrate a timing diagrams for selected signals of the soft-start circuit 108 illustrated in FIGS. 3 and 6. More particularly, FIG. 7a illustrates a timing diagram for the supply voltage V_(CC), while FIG. 7b illustrates a timing diagram for the signal V_(START1). Upon start-up, the supply voltage V_(CC) is zero or nearly zero. The supply voltage V_(S) is applied to the forward converter 200 (FIG. 4). In response, the supply voltage V_(CC) gradually ramps up, as shown in FIG. 7a, due to the capacitor C₄ (FIG. 4) charging through the bleed resistor R₃ (FIG. 4). As V_(CC) rises to a level sufficient to allow circuits of the controller 100 (FIG. 3) to begin operating, the output of the comparator CMP₅ (FIG. 3) holds the transistor M₂ (FIG. 6) off while the comparator CMP₆ (FIG. 6) holds the transistor M₃ (FIG. 6) off. In the preferred embodiment, V_(REF2) (FIG. 3) is 12 volts, while V_(REF4) (FIG. 6) is 13 volts. While V_(CC) is below V_(REF2), the signal V_(START1) is zero volts (any residual charge is discharged to the first ground node through the resistor R₅ (FIG. 6)). This is shown in FIG. 7b where the signal V _(START1) is substantially at ground level prior to the time t₁ where V_(CC) reaches V_(REF2) (e.g., 12 volts).

The PWM comparator CMP₃ (FIG. 3) compares the smaller of the signal V_(DC) (or V_(EA1)) and the signal V_(START1) to the signal I_(SENX) for controlling the duty cycle of the transistor M₁ (FIG. 4). FIG. 9 illustrates a schematic diagram of the PWM comparator CMP₃ illustrated in FIG. 3. Referring to FIGS. 3-4, upon start-up, the output voltage V_(OUT1) is zero or nearly zero. As such, the error signal V_(DC) (or V_(EA1)) is relatively large, whereas, the signal V_(START1) is substantially zero, as explained above. Accordingly, the output of the comparator CMP₃ prevents the transistor M₁ from turning on.

Eventually, the supply V_(CC) reaches a level above V_(REF2) (e.g., 12 volts). This is shown occurring at the time t₁ in FIG. 7a. In response, the output of the comparator CMP₅ (FIG. 3) transitions from a logic low voltage to a logic high voltage and the transistor M₂ (FIG. 6) is turned on. As a result, the current source I₁ (FIG. 6) generates a current which forms the voltage signal V_(START1) across the resistor R₅ (FIG. 6). The current generated by the current source I₁ is representative of the level of the supply V_(CC). In the preferred embodiment, the current generated by the current source I₁ is substantially proportional to the level of the supply V_(CC) once the supply voltage reaches the level of the reference voltage V_(REF2). Thus, the current I₁ can be given as: I₁=K₁*(V_(CC)−V_(REF2)), where K₁ is a constant. After V_(CC) surpasses the level of V_(REF2), then the signal V_(START1) increases as V_(CC) increases. This is shown in FIG. 7b by the signal V_(START1) gradually rising after the time t₁, as does the supply voltage V_(CC) in FIG. 7a.

After the time t₁, the output of the comparator CMP₃, no longer prevents the transistor M₁ (FIG. 4) from turning on. Rather, the signal V_(START1) is compared to the signal I_(SENX) by the comparator CMP₃ so as to allow the transistor M₁ to turn on for a portion of each cycle of the clock signal V_(CLK1). In response, V_(CC) increases further and the output voltage V_(OUT1) (FIG. 4) starts to increase. As the level of the supply V_(CC) continues to gradually increase, so does the level of the signal V_(START1). Accordingly, the duty cycle of the transistor M₁ gradually increases after the time t₁. In response, the output voltage V_(OUT), ramps up while the error signal V_(DC) (or V_(EA1) ) begins to fall.

Eventually, the error signal V_(DC) (or V_(EA1) ) can fall to a level below that of the signal V_(START1), at which point, the soft-start circuit 108 no longer affects operation of the forward converter 200. In the preferred embodiment, however, once the supply voltage V_(CC) rises to a level of V_(REF4) (e.g., 13 volts), the output of the comparator CMP₆ (FIG. 6) transitions from a logic low voltage to a logic high voltage. As a result, the transistor M₃ (FIG. 6) turns on. In response, the current source I₂ (FIG. 6) generates a current which serves to rapidly increase the signal V_(START1) to a level which is above the range of possible levels for the error signal V_(DC) (or V_(EA1) ). This is shown in FIG. 7a where the supply voltage V_(CC) exceeds V_(REF4) (e.g., 13 volts) at the time t₂ and, at the same time, the signal V_(START1) rapidly increases, as shown in FIG. 7b. Preferably, the signal V_(START1) is rapidly increased to a level of 5 volts because, in the preferred embodiment, the error signal V_(DC) and V_(EA1), are constrained between zero and five volts. Because V_(START1) is raised to a level higher than the error signal V_(DC), the duty cycle of the transistor M₁ is no longer affected by the soft-start circuit 108 (FIG. 3) and the forward converter 200 operates to regulate the output voltage V_(OUT1) at the desired level, as described above in reference to FIGS. 3-5.

In an alternate embodiment, the current source I₂ (FIG. 6) is omitted and, instead, the drain of the transistor M₃ is coupled to the supply V_(CC) or to an appropriate reference voltage. When the transistor M₃ is activated by the signal from the comparator CMP₆ (FIG. 6), the signal V_(START1) is rapidly increased to the level of the supply V_(CC) or to the level of the appropriate reference voltage. In response, the comparator CMP₃ compares the signal I_(SENSE1) to the error signal V_(DC) (or V_(EA1) ) rather than to the signal V_(START1).

During operation, the supply voltage V_(CC) is preferably maintained at a level of approximately 14 volts, as shown in FIG. 7a. The comparator CMP₆ (FIG. 6) preferably exhibits a hysteretic characteristic so as to prevent the soft-start circuit 108 from affecting the duty cycle of the transistor M₁ in the event that the supply voltage V_(CC) drops. Rather, the soft-start circuit 108 is preferably activated only upon start-up. If the supply voltage V_(CC) falls below a predetermined low level (e.g., 8 volts), then this indicates an error condition and the controller 100 is preferably shut down until it is reset.

A feature of the soft start circuit 108 is that the current source I₂ generates a current which is related to the supply voltage V_(CC), rather than generating a constant current. This enables the signal V_(START1), formed across the resistor R₅ (FIG. 6), to increase gradually and reliably although an external capacitor is not required for forming the signal V_(START1). This advantageously reduces the number pins required when the controller 100 as implemented as an integrated circuit. As shown in FIGS. 3-4, such an integrated circuit can be implemented having eight pins and no capacitor, external or otherwise, is required for forming the signal V_(START1).

FIG. 8 illustrates a more detailed schematic diagram of the soft-start circuit illustrated in FIGS. 3 and 6. The signal DCOK remains a logic high voltage until internal reference voltages of the controller 100 (FIG. 3) reach their operating levels and, then, transitions to a logic low voltage.

During operation of the forward converter 200 (FIG. 4) it is possible for the load 204 which receives the output voltage V_(OUT1) to draw a low level of current. In which case, the output voltage V_(OUT1) tends increase due to the transistor M₁ being switched on and off for each cycle of the clock signal V_(CLK1). In the preferred embodiment, the pulse skip circuit 102 (FIG. 3) disables switching of the transistor M₁ under certain light load conditions. The pulse skip circuit 102 preferably disables switching of the transistor M₁ by changing its output signal P_(SKIP) from a logic low voltage to a logic high voltage. In response, the output of the NOR gate U₃ (FIG. 3) changes from a logic high voltage to a logic low voltage. As a result, the output of the NOR gate U₆ (FIG. 3) and, thus, the signal PWM OUT, remain a logic low voltage, so long as the signal P_(SKIP) is a logic high voltage, regardless of a level of the output of the comparator CMP₃ (FIG. 3).

FIG. 10 illustrates a schematic block diagram of the pulse skip circuit 102 illustrated in FIG. 3. A first terminal of a current source 13 and a first terminal of a current source I₄ are coupled to receive the supply voltage V_(CC). A second terminal of the current source I₃ is coupled to a non-inverting input of a comparator CMP₇, to a first terminal of a resistor R₆, and to a collector of bipolar transistor Q₁. A second terminal of the current source I₄ is coupled to a collector of a bipolar transistor Q₂, to a base of the transistor Q₂ and to a base of the transistor Q₁. An emitter of the transistor Q₁, an emitter of the transistor Q₂ and a second terminal of the resistor R₆ are coupled to the first ground node. The error signal V_(DC) (or V_(EA1)) is coupled to an inverting input of the comparator CMP₇. An output of the comparator CMP₇ forms the signal P_(SKIP).

In the preferred embodiment, the current source I₃ generates a constant current while the current source I₄ generates a current which is related to the level of the supply V_(CC). The transistors Q₁ and Q₂ form a current mirror such that a current flowing through the transistor Q₁ is equal to the current generated by the current source I₄. Thus, a pulse skip reference voltage signal V_(SKIP) formed across the resistor R₆ is inversely related to the level of the supply V_(CC). In the preferred embodiment, the voltage signal V_(SKIP) varies substantially in reverse proportion to the level of the supply V_(CC). Accordingly, the voltage signal V_(SKIP) can be given as V_(SKIP) =K₂−(K₃*V_(CC)), where K₂ and K₃ are constants.

When the load 204 (FIG. 4) draws a sufficient level of current that the output voltage V_(OUT1) is regulated within a predetermined range, the output P_(SKIP) of the comparator CMP₇ is logic low voltage. This is because the level of the error signal V_(DC) (or V_(EA1) ) is higher than the signal V_(SKIP). However, when the output voltage V_(OUT1) rises, as tends to occur when the load 204 draws a low level of current, the error signal V_(DC) (or V_(EA1)) decreases. Assuming the level of the error signal V_(DC) (or V_(EA1) ) falls below the level of the signal V_(SKIP), then the transistor switch M₁ (FIG. 4) will be held off by the signal P_(SKIP) changing to a logic high voltage until the error signal V_(DC) (or V_(EA1)) rises again to level sufficient to cause the output of the comparator CMP₇ to return the signal P_(SKIP) to a logic low voltage. Note that when the transistor switch M₁ is disabled, then the error signal V_(DC) (or V_(EA1)) will tend to rise since no power is delivered to the load 204 when the transistor M₁ is disabled from switching.

When the level of the voltage source V_(S) increases, more energy is transferred by the transformer T₁ (FIG. 4) for each cycle of the transistor M₁ (FIG. 4) than when the voltage source V_(S) is at a lower level. It is expected that when the level of the voltage source V_(S) increases, the supply voltage V_(CC) also increases. Because the pulse skip reference voltage signal V_(SKIP) is preferably inversely related to the supply voltage V_(CC), when the supply voltage V_(CC) is at a higher level, the output voltage V_(OUT1) must rise to a higher level before the signal P_(SKIP) is asserted in comparison to when the supply voltage V_(CC) is at a lower level. Forming the signal V_(SKIP) such that it is inversely related to the supply voltage V_(CC) tends to aid in spreading out the pulses in the control signal PWM OUT under light load conditions. This tends to conserve supply power and produces less noise than if periodic bursts of several pulses appeared the switch control signal PWM OUT. In addition, the levels of the current sources I₃, I₄ and the value of the resistor R₆ are preferably selected so as to aid in evenly spreading out in time to the extent practical the individual pulses which appear in the switch control signal PWM OUT.

FIG. 11 illustrates a more detailed schematic diagram of the pulse skip circuit illustrated in FIGS. 3 and 10.

FIG. 12 illustrates a block schematic diagram of the current limit circuit 106 illustrated in FIG. 3. The signal I_(SENSE1) (FIG. 3) is coupled to a non-inverting input of a comparator CMP₈ and to an inverting input of a comparator CMP₉. An inverting input of the comparator CMP₈ is coupled to a reference voltage V_(REF5). A non-inverting input of the comparator CMP₉ is coupled to a reference voltage V_(REF6). An output of the comparator CMP₈ is coupled to a first input of a logic OR gate U₉. An output of the comparator CMP₉ is coupled to a second input of the OR gate U₉. An output of the OR gate U₉ forms the signal I_(LIM) (FIG. 3).

In the preferred embodiment, the reference voltage V_(REF5) is positive (e.g., 1.5 volts), and the reference voltage V_(REF6) is negative (e.g., −100 mV). Thus, when the signal I_(SENSE1) is within the range of −100 mV to 1.5 volts, then the signal I_(LIM) is a logic low voltage. As a result, the transistor M₁ (FIG. 4) is turned on and off, as explained above, to control the output voltage V_(OUT1). However, when the signal I_(SENSE1) is outside the range of −100 mV to 1.5 volts, this indicates an over-current condition. In response, the signal I_(LIM) transitions from a logic low voltage to a logic high voltage. This opens (turns off) the transistor M₁ such that its duty cycle falls to zero. The transistor M₁ is disabled so long as the over-current condition persists.

FIG. 13 illustrates a schematic diagram of a PWM post regulator 300 which incorporates the integrated circuit controller 100 illustrated in FIG. 3. A voltage source V_(S) is coupled to a first terminal of a resistor R₇ and to a first terminal of a primary winding of a transformer T₂. A second terminal of the primary winding of the transformer T₂ is coupled to a drain of an MOS transistor M₄. A source of the transistor M₄ is coupled to a first ground node. An output V_(CLK2) of an oscillator 302 is coupled to Pin 6 of the controller 100 and to an input of an inverter U₁₀. An output of the inverter U₁₀ forms a signal {overscore (V)}_(CLK) ₂ and is coupled to a gate of the transistor M₄.

A first terminal of a first secondary winding of the transformer T₂ is coupled to an anode of a diode D₄. A cathode of the diode D₄ is coupled to a drain of a MOS transistor M₅. A gate of the transistor M₅ is coupled to Pin 7 of the controller 100. A source of the MOS transistor M₅ is coupled to a cathode of a diode D₅ and to a first terminal of an inductor L₂. A second terminal of the inductor L₂ is coupled to a first terminal of a capacitor C₅, to a first output node, and to a first terminal of a resistor R_(DIV1). A second terminal of the resistor R_(DIV1) is coupled to a first terminal of a resistor R_(DIV2) and to Pin 1 of the integrated circuit controller 100. A second terminal of the resistor R_(DIV2) is coupled to a second ground node. An output voltage V_(OUT2) is formed at the first output node. A second terminal of the secondary winding of the transformer T₂ is coupled to an anode of the diode D₅, to a first terminal of a sensing resistor R_(SENSE2), and to a first terminal of a resistor R_(RAMP). A second terminal of the resistor R_(RAMP) is coupled to a first terminal of a capacitor C_(RAMP) and to Pin 3 of the controller 100. A second terminal of the capacitor C_(RAMP), a second terminal of the resistor R_(SENSE2) and a second terminal of the capacitor C₅ are coupled to the second ground node. A load 304 can be coupled across the capacitor C₅. A signal I_(SENSE2) is formed at the second terminal of the resistor R_(RAMP) and is representative of a level of current passing through the secondary winding of the transformer T₂ when the transistor M₅ is active.

Pin 2 of the controller 100 is coupled to a first terminal of a resistor R_(Z) and to a first terminal of a capacitor C_(P). A second terminal of the resistor R_(Z) is coupled to a first terminal of a capacitor C_(Z). A second terminal of the capacitor C_(P) and a second terminal of the capacitor C_(Z) are coupled to the second ground node. The elements R_(Z), C_(P) and C_(Z) are preferably included for performing compensation/filtering of the current sense ramp I_(SENSE2).

A first terminal of a second secondary winding of the transformer T₂ is coupled to an anode of a diode D₆. A cathode of the diode D₆ is coupled to a first terminal of a capacitor C₆, to a second output node, to a second terminal of the resistor R₇, to Pin 4 of the controller 100 and to Pin 5 of the controller 100. A supply voltage V_(CC) is formed at the second output node. A second terminal of the second secondary winding of the transformer T₂ and a second terminal of the capacitor C₆ are coupled to the second ground node.

As shown in FIG. 13, the output voltage V_(OUT2) is coupled to Pin 1 of the integrated circuit controller 100 via a resistive divider formed of the resistors R_(DIV1) and R_(DIV2), whereas, the optical isolator 202 illustrated in FIG. 4 is omitted from FIG. 13. Accordingly, the error signal V_(EA1) is active, whereas, the error signal V_(DC) is not active. The error signal V_(EA1), is representative of a difference between the output voltage V_(OUT2) and a desired level for the output voltage V_(OUT2). The error signal V_(EA1), is utilized by the comparator CMP₃ (FIG. 3) for regulating the output voltage V_(OUT2), as explained herein.

FIGS. 14a-d illustrate timing diagrams for selected signals of the post regulator 300 illustrated in FIG. 13. More particularly, FIG. 14a illustrates a timing diagram for the clock signal {overscore (V)}_(CLK2); FIG. 14b illustrates a timing diagram for the clock signal V_(CLK2); FIG. 14c illustrates a timing diagram for the signal PWM OUT; FIG. 14d illustrates a timing diagram for the signal I_(SENSE2); and FIG. 14e illustrates a timing diagram for the error signal V_(EA1) and a timing diagram for the absolute value of the sensing signal I_(SENSE2), i.e. |I_(SENSE)|.

Note that because the clock signal V_(CLK2) is applied to Pin 6 of the controller 100, the clock signal V_(CLK1) (FIG. 3) is synchronous with the clock signal V_(CLK2). As a result, the transistor M₅ (FIG. 13) is controlled synchronously with the transistor M₄. This ensures that the first secondary winding of the transformer T₂ is energized when the transistor M₅ is turned on.

In the preferred embodiment, the clock signals V_(CLK2) and {overscore (V)}_(CLK2) each have a fifty-percent (50%) duty cycle and are one-hundred-eighty degrees (180°) out of phase with each other, as illustrated in FIGS. 14a-b. When the signal V_(CLK2) is a logical high voltage, the signals {overscore (V)}_(CLK2) and PWM OUT are each a logical low voltage, as illustrated in FIGS. 14a-c. Formation of the signal PWM OUT is described in above with reference to FIGS. 3 and 5. Because the signals {overscore (V)}_(CLK2) and PWM OUT are each a logical low voltage, the transistors M₄ (FIG. 13) and M₅ (FIG. 13) are off. When the clock signal V_(CLK2) transitions to a logical low voltage, the signals {overscore (V)}_(CLK2) and PWM OUT each transition to a logical high voltage, as shown in FIGS. 14b-c. As a result, the transistors M₄ and M₅ are turned on. Because the transistors M₄ and M₅ are both on, current flows in the primary winding of the transformer T₂, which induces a current to flow in the secondary windings of the transformer T₂.

Current flowing in the first secondary winding of the transformer T₂ is drawn through the sensing resistor R_(SENSE2), thereby forming the voltage signal I_(SENSE2). Because the second terminal of the sensing resistor R_(SENSE2) is coupled to the ground node, the signal I_(SENSE2), formed at the first terminal of the sensing resistor R_(SENSE2) is negative in polarity, as shown in FIG. 14d. The current sense circuit 104 (FIG. 3) forms the signal I_(SENX) which is representative of the absolute value of the signal I_(SENSE2) As shown in FIG. 14e, the absolute value of the signal I_(SENSE2) begins to rise upon the clock signal V_(CLK) ₂ transitioning from a logical high voltage to a logical low voltage. When the absolute value of the signal I_(SENSE2) reaches a level of the error signal V_(EA1) as shown in FIG. 14e, then the output of the comparator CMP ₃ changes from a logic high voltage to a logic low voltage. In response, the signal PWM OUT changes from a logic high voltage to a logic low voltage, as shown in FIG. 14c. This turns the transistor M₅ off. This cycle repeats upon a next transition in the clock signal V_(CLK2).

In this manner, operation of the transistor M₅ is synchronized with operation of the transistor M₄ and a duty cycle for the transistor M₅ is controlled so as to regulate the output voltage V_(OUT2). Because the output voltage V_(OUT2) is regulated by controlling the duty cycle of the transistor M₅, which is located on the same side of the transformer T₂ as the load 304, the power converter illustrated in FIG. 13 is referred to as a post regulator. This is in contrast to the forward converter illustrated in FIG. 4 in which the transistor M₂ is located on the opposite side of the transformer T₁ from the load 204. An advantage of the controller 100 of the present invention is that it can be utilized for controlling switching in a forward converter or a post regulator without modification.

The above-described regulation of the output voltage V_(OUT2) can be altered or interrupted under certain conditions. For example, in the event of an overvoltage condition, the signal OVP (FIG. 3) transitions from a logic high voltage to a logic low voltage. In response, the AND gate U₈ (FIG. 3) holds the transistor M₅ (FIG. 13) off, thereby disabling switching of the transistor M₅. In addition, under start-up conditions, the soft-start circuit 108 (FIG. 3) ensures that the duty cycle for the transistor M₅ is gradually increased. Further, the pulse skip circuit 102 (FIG. 3) temporarily disables the transistor M₅ when the load 304 (FIG. 13) draws a low level of current. Also the current limit circuit 106 (FIG. 3) disables the transistor M₅ when the current through the secondary winding of the transformer T₂ becomes excessive.

As mentioned, a feature of the present invention allows the same integrated circuit controller 100 to be utilized in a PWM forward converter 200, as illustrated in FIG. 4, and in a PWM post regulator 300, as illustrated in FIG. 13, by modifying only circuitry external to the integrated circuit 100. An aspect of the invention which is in furtherance of this feature is the current sense circuit 104 (FIG. 3). As mentioned above, the current sense circuit 104 ensures that the signal I_(SENX), which is applied to the comparator CMP₃ (FIG. 3), is positive in polarity whether the signal I_(SENSE1) (FIG. 4), which is positive in polarity, or the signal I_(SENSE2) (FIG. 13), which is negative in polarity, is applied to the input of the current sense circuit 104.

FIG. 15 illustrates a block schematic diagram of the current sense circuit 104 illustrated in FIG. 3. The signal I_(SENSE1) (FIG. 4) or the signal I_(SENSE2) (FIG. 13) can be coupled to an input terminal of the current sense circuit 104. The input terminal is coupled to a non-inverting input of an amplifier U₁₁ and to a first terminal of a resistor R₈.

A first terminal of a current source I₅ and a first terminal of a current source I₆ are coupled to the supply voltage V_(CC). A second terminal of the resistor R₈ is coupled to an emitter of a bipolar transistor Q₃ and to a drain of a MOS transistor M₆. A base of the transistor Q₃ is coupled to a base of a bipolar transistor Q₄, to a collector of the transistor Q₃ and to a second terminal of the current source I₅. An emitter of the transistor Q₄ is coupled to a first terminal of a resistor R₉. A second terminal of the resistor R₉ is coupled to the ground node. A collector of the transistor Q₄ is coupled to a second terminal of the current source I₆ and to a base of the transistor M₆.

A drain of the transistor M₆ is coupled to a base of a bipolar transistor Q₆, to a base of a bipolar transistor Q₇, to a collector or the transistor Q₆ and to a collector of a bipolar transistor Q₅. An emitter of the transistor Q₆ and an emitter of the transistor Q₇ are coupled to the supply voltage V_(CC). An output of the amplifier U₁₁ is coupled to a base of the transistor Q₅. An emitter of the transistor Q₅ is coupled to an inverting input of the amplifier U₁₁ and to a first terminal of a resistor R₁₀. A collector of the transistor Q₇ is coupled to a first terminal of a resistor R₁₁. A second terminal of the resistor R₁₀ and a second terminal of the resistor R₁₁ are coupled to the ground node. The signal I_(SENX) is formed at the first terminal of the resistor R₁₁. In the preferred embodiment, the resistors R₈, R₉. R₁₀ and R₁₁ are equal in value. In addition, the current sources I₅ and I₆ preferably provide equal currents.

The signal I_(SENSE1) is preferably a positive value which is normally within a range of zero to 1.5 volts. When a positive voltage signal, such as the signal I_(SENSE1), is coupled to the input terminal of the current sense circuit 104 the amplifier U₁₁ turns on the transistor Q₅. In addition, the transistor M₆ is off due to its gate-to-source voltage being negative. A voltage formed at the first terminal of the resistor R₁₀ is substantially proportional to the signal I_(SENSE1). A current through the resistor R₁₀ is, therefore, also substantially proportional to the signal I_(SENSE1) and is mirrored by the transistors Q₆ and Q₇ such that a current which flows through the transistor Q₇ and the resistor R₁₁ is substantially proportional to the signal I_(SENSE1). Because the current through the resistor R₁₁ is substantially proportional to the signal I_(SENSE), the voltage signal I_(SENX) which is formed at the first terminal of the resistor R₁₀ is also substantially proportional to the signal I_(SENSE1).

When a signal of zero volts is applied to the input terminal of the current sense circuit 104, the transistor Q₅ is turned off by the amplifier U₁₁. In addition, the transistor M₆ is substantially off due to its gate-to-source voltage being less than is required to turn on the transistor M₆. Accordingly, the output signal I_(SENX) is also zero volts. An offset voltage, e.g., 100 mV, can be inserted in series with the non-inverting input of the amplifier U₁₁ to ensure that, under such conditions, the transistor Q₅ is off.

The signal I_(SENSE2) is preferably a negative value which is normally within the range of −100 mV to zero volts. When a negative voltage signal, such as the signal I_(SENSE2), is applied to the input terminal of the current sense circuit 104, the amplifier U₁₁ holds the transistor Q₅ off. In addition, the transistor M₆ is turned on as its gate-to-source voltage is pulled down by the signal I_(SENSE2). A voltage at the drain of the transistor M₆ is substantially equal to the signal I_(SENSE2). This turns on the transistors Q₆ and Q₇. The transistors Q₆ and Q₇ form a current mirror such that each draws a current from V_(CC) which is substantially proportional to the signal I_(SENSE2). The current through the transistor Q₇ forms the output signal I_(SENX) across the resistor R₁₁ as a positive value which is substantially proportional to the input signal I_(SENSE2).

Accordingly, the signal I_(SENX) is representative of the absolute value of the signal I_(SENSE1) or I_(SENSE2) applied to the input of the current sense circuit 104.

According to yet another aspect of the present invention, one or more additional post regulators 300, as shown in FIG. 13, can be coupled to respective secondary windings of a single transformer. FIG. 16 illustrates a power converter circuit in accordance with the present invention including multiple post-regulators coupled to respective windings of a single transformer. As illustrated in FIG. 16, a voltage source V_(S) is coupled to a first terminal of a primary winding of a transformer T₃. A second terminal of the primary winding of the transformer T₃ is coupled to a drain of a MOS transistor M₅. A source of the transistor M₅ is coupled to a first ground node. An output V_(CLK3) of an oscillator 400 is coupled to an input of an inverter U₁₂ and to each of three controllers 100A, 100B and 100C. An output of the inverter U₁₂ forms a clock signal {overscore (V)}_(CLK3) which is coupled to control the gate of the transistor M₅. Preferably, the clock signals V_(CLK3) and {overscore (V)}_(CLK3) each have a fifty-percent (50%) duty cycle and are one-hundred-eighty degrees (180°) out of phase with each other.

Each controller 100A, 100B, and 100C can be identical to the controller 100 illustrated in FIG. 3. A first secondary winding of the transformer T₃ is coupled to a first post-regulator 402 which is controlled by the controller 100A in a manner identical to the post regulator 300 illustrated in FIG. 13 for forming an output voltage V_(OUTA). A second secondary winding of the transformer T₃ is coupled to a second post-regulator 404 which is controlled by the controller 100B in a manner identical to the post regulator 300 illustrated in FIG. 13 for forming an output voltage V_(OUTB) . A third secondary winding of the transformer T₃ is coupled to a third post-regulator 406 which is controlled by the controller 100B in a manner identical to the post regulator 300 illustrated in FIG. 13 for forming an output voltage V_(OUTC). Each of the output voltages V_(OUTA), V_(OUTB), and V_(OUTC) can be regulated at a different level depending upon the requirements of the respective loads. While three post regulators 402, 404, and 406 are shown, it will be apparent that the present invention can be practiced with another number.

In an alternate embodiment, the clock signal supplied to the SYNC input of the controller 100 (or controllers 100A-c) is supplied by a prior power converter stage. For example, the prior power converter stage can be a power factor correction (PFC) converter. A suitable PFC converter is described in the related parent application of which this application is a continuation-in-part.

The present invention has been described in terms of specific embodiments incorporating details to facilitate the understanding of the principles of construction and operation of the invention. Such reference herein to specific embodiments and details thereof is not intended to limit the scope of the claims appended hereto. It will be apparent to those skilled in the art that modifications may be made in the embodiment chosen for illustration without departing from the spirit and scope of the invention. Specifically, it will be apparent to one of ordinary skill in the art that the device of the present invention could be implemented in several different ways and the apparatus disclosed above is only illustrative of the preferred embodiment of the invention and is in no way a limitation. For example, it would be within the scope of the invention to vary the values of the various components, current levels, and voltage levels disclosed herein. 

What is claimed is:
 1. A power converter having a magnetic element and a switch for selectively interrupting a current through the magnetic element wherein a controller for the switch comprises: a. current sensing means for sensing the current through the magnetic element wherein the current sensing means receives a first voltage signal representative of the current through the magnetic element and wherein the current sensing means includes means for forming a second voltage signal representative of an absolute value of the first voltage signal whereby the first voltage signal can be positive or negative; b. voltage sensing means for sensing an output voltage of the power converter; and c. control means coupled to the first sensing means and to the second sensing means for controlling a duty cycle of the switch based upon the second voltage signal and the output voltage.
 2. The power converter according to claim 1 wherein the magnetic element is a transformer having a primary winding and a secondary winding wherein the current through the magnetic element flows through a primary winding of the transformer and an induced current in the secondary winding is rectified for forming the output voltage.
 3. The power converter according to claim 1 wherein the magnetic element is a transformer having a primary winding and a secondary winding wherein the current through the magnetic element flows through a secondary winding of the transformer and is induced by a current in the primary winding.
 4. The power converter according to claim 3 wherein the transformer comprises one or more additional secondary windings and wherein the power converter further comprises one or more additional switches, one coupled to each additional secondary winding and one or more additional controllers, one coupled to control each switch based upon a current through the respective secondary winding thereby forming one or more additional output voltages, one for each additional secondary winding.
 5. The power converter according to claim 1 wherein the magnetic element is a transformer having a primary winding and a secondary winding wherein when the first voltage signal is positive, the current through the magnetic element flows through a primary winding of the transformer and an induced current in the secondary winding is rectified for forming the output voltage, and wherein when the first voltage signal is negative, the current through the magnetic element flows through a secondary winding of the transformer and is induced by a current through a primary winding of the transformer.
 6. The power converter according to claim 1 wherein the first voltage signal is formed across a resistor coupled in series with the magnetic element.
 7. The power converter according to claim I further comprising current limit means for opening the switch when the first voltage signal rises above a first limit wherein the first limit is positive and opening the switch when the first voltage signal falls below a second limit wherein the second limit is negative.
 8. The power converter according to claim 6 wherein the resistor has a first terminal and a second terminal wherein the first voltage signal is formed at the first terminal and the second terminal is coupled to a ground node.
 9. The power converter according to claim 8 wherein the first voltage signal is positive when the current through the magnetic element flows to the ground node and wherein the first voltage signal is negative when the current through the magnetic element flows from the ground node.
 10. The power converter according to claim 9 wherein the magnetic element is a transformer having a primary winding and a secondary winding wherein when the first voltage signal is positive, the current through the magnetic element flows through a primary winding of the transformer and an induced current in the secondary winding is rectified for forming the output voltage, and wherein when the first voltage signal is negative, the current through the magnetic element flows through a secondary winding of the transformer and is induced by a current through a primary winding of the transformer.
 11. The power converter according to claim 10 wherein the controller is an integrated circuit having no more than eight pins.
 12. A power converter having a magnetic element and a switch for selectively interrupting a current through the magnetic element wherein the power converter is selected from a group consisting of a forward converter and a post regulator, wherein an integrated circuit controller for the switch is configurable for the forward converter and for the post regulator without modification to the integrated circuit and wherein the integrated circuit comprises: a. a current sensor for sensing the current through the magnetic element wherein the magnetic element is a primary winding of a transformer when the power converter is a forward converter and wherein the magnetic element is a secondary winding of the transformer when the power converter is a post-regulator; b. voltage sensor for sensing an output voltage of the power converter; and c. a switch control circuit coupled to the current sensor and to the voltage sensor for controlling a duty cycle of the switch based upon the current through the magnetic element and the output voltage.
 13. The power converter according to claim 12 wherein the current sensor receives a first voltage signal representative of the current through the magnetic element and wherein the first voltage signal is positive when the magnetic element is the primary winding and wherein the first voltage signal is negative when the magnetic element is the secondary winding.
 14. The power converter according to claim 13 wherein the current sensor includes means for forming a second voltage signal representative of an absolute value of the first voltage signal.
 15. The power converter according to claim 12 wherein the integrated circuit controller has no more than eight pins. 